Method and apparatus to compute a noise power estimate in a WCDMA network

ABSTRACT

Method and apparatus for computing a noise power estimate in a wideband CDMA (WCDMA) network are disclosed and may include calculating a noise power estimate for a downlink channel based on an orthogonal sequence generated for a transmitted signal. The orthogonal sequence may be generated based on a slot number of the transmitted signal and/or a transmit diversity mode used for the transmitted signal. A portion of a plurality of dedicated physical channel (DPCH) pilot bits for the downlink channel may be summed to generate an in-phase (I) component and a quadrature (Q) component. The generated I component and the generated Q component may be multiplied by the orthogonal sequence to generate at least one noise I component and at least one noise Q component.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. Non-Provisional applicationSer. No. 11/462,547, filed Aug. 4, 2006, which is hereby expresslyincorporated by reference in its entirety.

This application also makes reference to:

-   U.S. application Ser. No. 11/355,110, filed on Feb. 15, 2006;-   U.S. application Ser. No. 11/355,222, filed on Feb. 15, 2006;-   U.S. application Ser. No. 11/355,109, filed on Feb. 15, 2006;-   U.S. application Ser. No. 11/355,111, filed on Feb. 15, 2006; and-   U.S. application Ser. No. 11/422,689, filed on Jun. 7, 2006.

Each of the above state applications is hereby incorporated herein byreference in its entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to wireless communication.More specifically, certain embodiments of the invention relate to amethod and apparatus for computing a noise power estimate in a widebandCDMA (WCDMA) network.

BACKGROUND OF THE INVENTION

Mobile communications has changed the way people communicate and mobilephones have been transformed from a luxury item to an essential part ofevery day life. The use of mobile phones is today dictated by socialsituations, rather than hampered by location or technology. While voiceconnections fulfill the basic need to communicate, and mobile voiceconnections continue to filter even further into the fabric of every daylife, the mobile Internet is the next step in the mobile communicationrevolution. The mobile Internet is poised to become a common source ofeveryday information, and easy, versatile mobile access to this datawill be taken for granted.

Third generation (3G) cellular networks have been specifically designedto fulfill these future demands of the mobile Internet. As theseservices grow in popularity and usage, factors such as cost efficientoptimization of network capacity and quality of service (QoS) willbecome even more essential to cellular operators than it is today. Thesefactors may be achieved with careful network planning and operation,improvements in transmission methods, and advances in receivertechniques. To this end, carriers need technologies that will allow themto increase downlink throughput and, in turn, offer advanced QoScapabilities and speeds that rival those delivered by cable modem and/orDSL service providers. In this regard, networks based on wideband CDMA(WCDMA) technology may make the delivery of data to end users a morefeasible option for today's wireless carriers.

In the case of a WCDMA downlink, multiple access interference (MAI) mayresult from inter-cell and intracell interference. The signals fromneighboring base stations compose intercell interference, which ischaracterized by scrambling codes, channels and angles of arrivalsdifferent from the desired base station signal. Spatial equalization maybe utilized to suppress inter-cell interference. In a synchronousdownlink application, employing orthogonal spreading codes, intra-cellinterference may be caused by multipath propagation. Due to the non-zerocross-correlation between spreading sequences with arbitrary timeshifts, there is interference between propagation paths afterdespreading, causing MAI. The level of intra-cell interference dependsstrongly on the channel response. In nearly flat fading channels, thephysical channels remain almost completely orthogonal and intra-cellinterference does not have any significant impact on the receiverperformance. Frequency selectivity is common for the channels in WCDMAnetworks.

Mobile networks allow users to access services while on the move,thereby giving end users freedom in terms of mobility. However, thisfreedom does bring uncertainties to mobile systems. The mobility of theend users causes dynamic variations both in the link quality and theinterference level, sometimes requiring that a particular user changeits serving base station. This process is known as handover (HO).Handover is the essential component for dealing with the mobility of endusers. It guarantees the continuity of the wireless services when themobile user moves across cellular boundaries.

WCDMA networks may allow a mobile handset to communicate with a multiplenumber of cell sites. This may take place, for example, for asoft-handoff from one cell site to another. Soft-handoffs may involvecell sites that use the same frequency bandwidth. On occasions, theremay be handoffs from one cell site to another where the two cell sitesuse different frequencies. In these cases, the mobile handset may needto tune to the frequency of the new cell site. Additional circuitry maybe required to handle communication over a second frequency of thesecond cell site while still using the first frequency for communicatingwith the first cell site. The additional circuitry may be an undesirableextra cost for the mobile handset. In addition, the mobile handset mayrequire different transmit power to establish and maintain acommunication link with the new cell site. In a handoff scenario, themobile handset may still be receiving a strong signal from the currentcell site and a weaker signal from the new cell site. In this regard,transmit power may have to be adjusted so that the handoff may beachieved and the mobile handset may begin to communicate with the newcell site. Conventional methods of calculating noise power utilize meanand/or variance, which results in a biased estimate of the noise power.Such biased estimates are often times inaccurate.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or apparatus for computing a noise power estimate in awideband CDMA (WCDMA) network, substantially as shown in and/ordescribed in connection with at least one of the figures, as set forthmore completely in the claims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is an exemplary diagram illustrating a WCDMA handsetcommunicating with two WCDMA base stations, in accordance with anembodiment of the invention.

FIG. 1B is a block diagram of an exemplary radio frame format of adownlink dedicated physical channel (DPCH), in accordance with anembodiment of the invention.

FIG. 2A is a block diagram illustrating determination of a noise powerestimate in a WCDMA network, which may be used in accordance with anembodiment of the invention.

FIG. 2B is a block diagram illustrating determination of a noise powerestimate in a WCDMA network utilizing an orthogonal sequence, inaccordance with an embodiment of the invention.

FIG. 3 is a flowchart illustrating exemplary steps for determining anoise power estimate in a WCDMA network, in accordance with anembodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method andapparatus for computing a noise power estimate in a wideband CDMA(WCDMA) network, and may include calculating a noise power estimate fora downlink channel based on an orthogonal sequence generated for atransmitted signal. The orthogonal sequence may be generated based on aslot number of the transmitted signal and/or a transmit diversity modeused for the transmitted signal. A portion of a plurality of dedicatedphysical channel (DPCH) pilot bits for the downlink channel may besummed to generate an in-phase (I) component and a quadrature (Q)component. The generated I component and the generated Q component maybe multiplied by the orthogonal sequence to generate one or morecorresponding I and Q noise components. The generated corresponding Iand Q noise components may be summed and the resulting I and Q summedcomponents squared, to yield corresponding I and Q noise components. Thesquared I and Q noise components may be summed and then normalized toyield a noise power estimate for the downlink channel. The normalizingmay comprise dividing the generated summed noise component by a numberof dedicated pilot bits per slot of the downlink channel.

FIG. 1A is an exemplary diagram illustrating a WCDMA handsetcommunicating with two WCDMA base stations, in accordance with anembodiment of the invention. Referring to FIG. 1A, there is shown amobile handset or user equipment 120, a plurality of base stations BS122 and BS 124, and a plurality of radio links (RL), RL₁ and RL₂coupling the user equipment 120 with the base stations BS 122 and BS124, respectively. The user equipment 120 may comprise a processor 142,a memory 144, and a radio 146. The radio 146 may comprise a transceiver(Tx/Rx) 147.

In accordance with an embodiment of the invention, methods forprocessing a plurality of dedicated physical channel (DPCH) pilot bitsfor the downlink channel disclosed herein may apply to diversity andnon-diversity wireless systems. Diversity wireless systems may comprisespace-time transmit diversity (STTD), closed loop 1 (CL1), and closedloop 2 (CL2) wireless systems.

Uplink power control (PC) is of paramount importance for CDMA-basedsystems because the capacity of such a system is a function of theinterference level. The power transmitted by all active user equipments(UE) within a network may be controlled to limit interference levels andalleviate well-known problems such as the “near-far” effect. If there ismore than one user active, the transmitted power of non-reference usersis suppressed by a factor which may depend on the partialcross-correlation between the code of the reference user and the code ofthe non-reference user. However, when a non-reference user is closer tothe receiver than the reference user, it is possible that theinterference caused by this non-reference user has more power than thereference user also referred to as the “near-far” effect. A UE mayutilize open-loop power-control to measure its received signal power andadjusts its transmit power accordingly. An active radio link (RL) mayutilize closed-loop power-control to measure the received signal powerfrom all user equipments and command individual user equipments to raiseor lower their transmit uplink power such that the receivedsignal-to-noise ratio (SNR) from all user equipments at the radio linksis the same.

The processor 142 may communicate and/or control a plurality of bitsto/from the base stations BS 122 and BS 124. The memory 144 may comprisesuitable logic, circuitry, and/or code that may store data and/orcontrol information. The radio 146 may comprise transmit circuitryand/or receive circuitry that may be enabled to calculate asignal-to-noise ratio (SNR) and/or a noise power estimate of a downlinkdedicated physical channel (DPCH) based on a plurality of transmit powercontrol (TPC) bits and/or a plurality of dedicated pilot bits receivedvia the downlink dedicated physical channel (DPCH), where the pluralityof TPC bits may not be known when they are received. The radio linksthat belong to the same radio link set may broadcast the same values oftransmit power control (TPC) bits. The radio links that belong todifferent radio link sets may broadcast different TPC bits. The userequipment 120 may receive TPC bits via multiple radio links, forexample, RL₁ and RL₂ simultaneously. During handover, the user equipment120 may simultaneously receive signals from multiple radio link sets.

The WCDMA specification defines the physical random access channel(PRACH) for mobile phone uplinks and the acquisition indicator channel(AICH) for BTS downlinks. Communication is established when the userequipment 120 completes its search for a base station, for example, BS122 and synchronizes its PRACH uplink signal with the BTS AICH downlinksignal. The base station may recognize a resulting PRACH preamble fromthe user equipment 120 and responds with an AICH to establish acommunication link. The user equipment 120 may use the PRACH to transmitits setting of its open loop power control to the base station 122.Incorrect data in the PRACH preamble or problems with the signal qualitymay cause missed connections, disrupt the capacity of the cell and/orprevent response from the base station 122.

FIG. 1B is a block diagram of an exemplary radio frame format of adownlink dedicated physical channel (DPCH), in accordance with anembodiment of the invention. Referring to FIG. 1B, there is shown aradio frame format 102, with a time period T_(f) equal to 10 ms, forexample. The radio frame 102 may comprise a plurality of slots, forexample, 15 slots. Each of the slots in the radio frame 102, forexample, slot # i 104 may comprise a plurality of dedicated physicaldata channels (DPDCH) and a plurality of dedicated physical controlchannels (DPCCH). The time period of each slot in the radio frame 102,for example, time period of slot # i may be equal to 10*2^(k) bits,where k=0 . . . 7, for example.

The DPDCH is a type of downlink channel, which may be represented as anI/Q code multiplexed within each radio frame 102. The downlink DPDCH maybe utilized to carry data, for example, data 1 154 comprising N_(data1)bits and data 2 160 comprising N_(data2) bits. There may be zero, one,or a plurality of downlink dedicated physical data channels on eachradio link.

The DPCCH is a type of downlink channel, which may be represented as anI/Q code multiplexed within each radio frame 102. The downlink DPCCH maybe utilized to carry control information generated at the physicallayer. The control information may comprise a transmit power control(TPC) block 156 comprising N_(TPC) bits per slot, a transport formatcombination indicator (TFCI) block 158 comprising N_(TFCI) bits per slotand a pilot block 162 comprising N_(pilot) bits per slot.

The pilot bits 162 are known a priori, that is, they are known whenreceived by a receiver. The term “a priori” means “formed or conceivedbeforehand.” Therefore, conventional methods of computing asignal-to-noise ratio (SNR) metric are based on multiplying the receivedsignal by a known sequence and thereafter computing a mean and varianceof the received signal. These conventional methods, however, result in abiased estimate of the noise power. Such biased estimates are oftentimes inaccurate.

In an embodiment of the invention, the quality and/or the noise power ofthe downlink control channel transmitted with the downlink dedicatedphysical channel (DPCH) may be determined. Within one downlink DPCH,dedicated data may be transmitted in a time-multiplex manner withcontrol information. The control information may comprise pilot bits,transport format combination indicator (TFCI) bits and/or transmit powercontrol (TPC) bits.

FIG. 2A is a block diagram illustrating determination of a noise powerestimate in a WCDMA network, which may be used in accordance with anembodiment of the invention. Referring to FIG. 2A, there is shown aplurality of pilot extraction fingers for a given radio link (RL), forexample, pilot extraction finger i 202 a through pilot extraction fingerj 204 a, summing blocks 206 a, 211 a, . . . , 216 a, 226 a, 232 a, and236 a. There is also shown multiplying blocks 207 a, 209 a, 230 a,squaring blocks 218 a, 220 a, 228 a, and divider blocks 222 a, 224 a,and 234 a.

In a multipath-fading environment, a receiver structure, such as thestructure illustrated in FIG. 2A, may assign fingers to the multiplereceived paths, for example, pilot extraction finger i 202 a and pilotextraction finger j 204 a. Those fingers belonging to the same radiolink (RL) set may be summed by the summing block 206 a to generate pilotI sum 208 a and pilot Q sum 210 a. A noise power estimate may becomputed utilizing the pilot bits.

Within a wireless communication network, a received wireless signal maybe modeled as a stationary random variable. From statistical theory, itfollows that if x is a random variable, then its variance σ_(x) may becomputed utilizing the following equation:σ_(x) ² =E[(x−E[x])² ]=E[x ²]−(E[x])²  (1.)Consequently, the noise power of a received signal may be computed byestimating the variance of the received signal.

In operation, pilot bits from all fingers 202 a, . . . , 204 a may besummed by the summing block 206 a. The resulting pilot Q sum 210 a andthe pilot I sum 208 a may be communicated to the multiplying blocks 209a and 207 a, respectively. In circumstances when non-diversity flatfading is present, the soft value of each of the dedicated pilot bits ateach slot may be obtained from the receiving user equipment utilizing adedicated DPCH control message. The i-th pilot symbol may be representedby the following equation:

$\begin{matrix}{{z_{i} = {{\sqrt{\frac{S_{DED}}{2}}( {{ISeq} + {jQSeq}} ){h}^{2}} + {n_{i}h^{*}}}},} & (2.)\end{matrix}$where S_(DED) comprises the transmit signal power, ISeq and QSeqcomprise transmitted I and Q sequences, Iseq+jQSeq comprises a pilotsymbol, h comprises the channel gain at the finger i, and n_(I)comprises the noise of the signal received at the user equipment. Thenumber of dedicated pilot bits per slot may be denoted by num_ded, whichmay result in num_ded/2 pilot symbols per slot.

Referring to FIG. 2A, the pilot I sum 208 a and the pilot Q sum 210 acommunicated from the summing block 206 a, may be expressed using thefollowing equations:

$\begin{matrix}{{I = {{\sqrt{\frac{S_{DED}}{2}}{ISeq}{h}^{2}} + {{{Re}( {n_{i}h^{*}} )}\mspace{14mu}{and}}}}{Q = {{\sqrt{\frac{S_{DED}}{2}}{QSeq}{h}^{2}} + {{Im}( {n_{i}h^{*}} )}}}} & (3.)\end{matrix}$After the pilot I sum 208 a and the pilot Q sum 210 a are generated bythe summing block 206 a, each pilot sum 208 a and 210 a may bemultiplied by its own sequence I Seq 238 a and Q Seq 240 a,respectively, to remove phase rotation, using the multiplying blocks 207a and 209 a. The I and Q post-derotation components may be expressedusing the following equations:

$\begin{matrix}{{{I = {{{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + {{{Re}( {n_{i}h^{*}} )}{ISeq}}} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + n_{I}}}},{and}}{{Q = {{{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + {{{Im}( {n_{i}h^{*}} )}{QSeq}}} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + n_{Q}}}},}} & (4.)\end{matrix}$where n_(I) and n_(Q) comprise the noise components on the I and Qbranches, for which power may be estimated.

In one embodiment of the invention, the noise power estimate may becalculated using the subtraction of the mean of the received signal tothe square (E[x])² from the mean of the received signal power E[x²]. TheI and Q post-derotation components may be summed by the summing blocks211 a and 212 a, respectively, over num_ded/2. The summed I and Qpost-derotation components may then be normalized by the divider blocks222 a and 224 a, respectively. For example, the divider blocks 222 a and224 a may divide the summed I and Q post-derotation components bynum_ded/2 to achieve normalization. The outputs of the divider blocks222 a and 224 a may be denoted as mean) 242 a and meanQ 244 a and may beexpressed using the following equations:

$\begin{matrix}{{{meanI} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + {\frac{1}{{num\_ ded}/2}{\sum\limits_{i}^{\;}{{n_{I}(i)}\mspace{14mu}{and}}}}}}{{meanQ} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + {\frac{1}{{num\_ ded}/2}{\sum\limits_{i}^{\;}{n_{Q}(i)}}}}}} & (5.)\end{matrix}$

Referring back to equation (1), the mean of the received signal to thesquare (E[x])² may be computed using the following equation:(E[x])² =K·(meanI+meanQ)²,  (6.)where K may comprise a scaling factor, which may be based on a hardwareconstant. The computation of (meanI+meanQ)² may be performed by thesumming block 226 a and the squaring block 228 a, and the multiplicationby the scaling factor K may be performed by the multiplying block 230 a.

The I and Q post-derotation components may be squared by the squaringblocks 218 a and 220 a, respectively. The squared components may beexpressed by the following equations:

$\begin{matrix}{{{I^{2} = {( {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + n_{I}} )^{2}\mspace{14mu}{and}}}{Q^{2} = ( {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}} + n_{Q}} )^{2}}}\mspace{11mu}} & (7.)\end{matrix}$The squared components I² and Q² may be summed by the summing blocks 214a and 216 a, respectively, over num_ded/2. The summed squared componentsmay then be combined by the summing block 232 a. The combined output ofthe summing block 232 a may be normalized by the divider block 234 a.For example, the divider block 234 a may divide the combined output ofthe summing block 232 a by num_ded to achieve normalization, therebygenerating the mean meansq 246 a of the received signal power E[x²]. Themeans 246 a may be expressed using the following equation:

$\begin{matrix}{{meansq} = {\frac{1}{num\_ ded}( {{\sum\limits_{i = 1}^{{{num}\_{ded}}/2}{I^{2}(i)}} + {\sum\limits_{i = 1}^{{{num}\_{ded}}/2}{Q^{2}(i)}}} )}} & (8.)\end{matrix}$

Referring again to equation (1), the mean of the received signal powerE[²] may be computed using the following equation:E[x²]=meansq  (9.)In addition, it may follow from equation (1) that noise power may becalculated by subtracting the mean of the received signal to the square(E[x])² from the mean of the received signal power E[x²], using thesumming block 236 a. In this regard, the noise power estimate Npilot 248a may be calculated using the following equation:Npilot=meansq−K·(meanI+meanQ ²  (10.)

FIG. 2B is a block diagram illustrating determination of a noise powerestimate in a WCDMA network utilizing an orthogonal sequence, inaccordance with an embodiment of the invention. Referring to FIG. 2B,there is shown a plurality of pilot extraction fingers for a given radiolink (RL), for example, pilot extraction finger i 202 b through pilotextraction finger j 204 b, summing blocks 206 b, 216 b, and 222 b. Thereis also shown a loading block 214 b, a multiplying block 212 b, squaringblocks 218 b, 220 b, and a divider block 224 b.

In a multipath-fading environment, a receiver structure, such as thestructure illustrated in FIG. 2B, may assign fingers to the multiplereceived paths, for example, pilot extraction finger i 202 b at d pilotextraction finger ┘204 b. Those fingers belonging to the same radio link(RL) set may be summed by the summing block 206 b to generate pilot Isum 208 b and pilot Q sum 210 b. In one embodiment of the invention, anoise power estimate may be computed utilizing the pilot bits and one ormore orthogonal sequences generated using a transmitted sequence ofsymbols. In a non-diversity flat fading case, the soft value of each ofthe dedicated pilot bits at each slot of a downlink channel may beobtained from the hardware and the i-th pilot symbol may be representedby the following equation:

$\begin{matrix}{{z_{i} = {{\sqrt{\frac{S_{DED}}{2}}x_{i}{h}^{2}} + {n_{i}h^{*}}}}\mspace{11mu}} & (11.)\end{matrix}$

The number of dedicated pilot bits per slot may be denoted by num_dedand num_ded/2 dedicated pilot symbols may be stacked in a vectoraccording to the following equation:

$\begin{matrix}{{\underset{\_}{z} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}\underset{\_}{x}} + {\underset{\_}{n}}^{\prime}}},} & (12.)\end{matrix}$where n′ may be the post-combining noise, the power of which may beestimated. The pilot symbol sequence

${\underset{\_}{x}}^{T} = \lbrack {x_{0},x_{1},x_{2},\ldots\mspace{14mu},x_{\frac{{num}\_{ded}}{2}1}} \rbrack$may be known before a sequence is received. In this regard, anorthogonal sequence

${\underset{\_}{y}}^{T} = \lbrack {y_{0},y_{1},y_{2},\ldots\mspace{14mu},y_{\frac{{num}\;\_\;{ded}}{2} - 1}} \rbrack$may be calculated such thaty ^(H) x=0.  (13.)Since the pilot symbols comprise −1s and 1s, the sequence in y may alsocomprise −1s and 1s.

Multiplying the received symbols z by y ^(H) may result in a sign changemanipulation on the received I and Q, so that the following equation maybe satisfied:y ^(H) z=y ^(H) n ^(′)  (14.)The variance of the noise component n′ may be expressed by the followingequation:

$\begin{matrix}{{\sigma_{n^{\prime}}^{2} = {{{h}^{2}I_{oc}} = {E\lbrack {n_{i}^{\prime}n_{i}^{\prime*}} \rbrack}}},{i = 0},\ldots\mspace{14mu},{\frac{num\_ ded}{2} - 1.}} & (15.)\end{matrix}$Where I_(oc) may be the power spectral density of a band limited whitenoise source (simulating interference from cells) as measured at the UEantenna connector. In instances when the orthogonal sequence y may benormalized, the following equation may be satisfied:y ^(H) y=1.  (16.)In this regard, the variance of y ^(H)n^(′) may be expressed by thefollowing equation:{circumflex over (N)} _(pilot) =E[y ^(H) n ^(′) n ^(′H) y]=σ _(n) _(′) ²=|h| ² I _(oc)  (17.)

Referring to FIG. 2B, the loading block 214 b may determine one or moreorthogonal sequences, such as the orthogonal sequences 228 b and 230 b,based on a slot number information 226 b and transmit diversity modeinformation 227 b. For example, different orthogonal sequences may begenerated for closed loop 1 (CL1) transmit diversity mode. The generatedone or more orthogonal sequences may be communicated to the multiplyingblock 212 b. The generated I component 208 b and the generated Qcomponent 210 b may be multiplied by the one or more orthogonalsequences, such as the orthogonal sequences 228 b and 230 b, to removethe signal component and generate at least one noise I component 232 band at least one noise Q component 234 b.

The at least one noise I component 232 b and the at least one noise Qcomponent 234 b may be summed by the summing block 216 b to generate atleast one summed noise I component 236 b and at least one summed noise Qcomponent 238 b. The at least one summed noise I component 236 b and theat least one summed noise Q component 238 b may be squared by thesquaring blocks 218 b and 220 b to generate at least one squared noise Icomponent and at least one squared noise Q component. The at least onesquared noise I component and the at least one squared noise Q componentmay be summed by the summing block 222 b to generate a summed noisecomponent. The summed noise component may then be normalized by thedivider block 224 b to determine a noise power estimate Npilot 240 b forthe downlink channel.

In a flat fading STTD environment with the number of pilot bits greaterthan 2, the soft value of each dedicated pilot bits at each slot may beobtained from the hardware. The i-th received dedicated pilot symbol forantenna 1 may be equal to:

$\begin{matrix}{z_{1i} = {{\sqrt{\frac{S_{DED}}{4}}( {{x_{1i}h_{1}} + {x_{2i}h_{2}}} )h_{1}^{*}} + {n_{i}h_{i}^{*}}}} & (18.)\end{matrix}$Similarly, for antenna 2,

$\begin{matrix}{z_{2i} = {{\sqrt{\frac{S_{DED}}{4}}( {{x_{1i}h_{1}} + {x_{2i}h_{2}}} )h_{2}^{*}} + {n_{i}h_{2}^{*}}}} & (19.)\end{matrix}$The set of num_ded/2 dedicated pilot symbols may be stacked in a vectoraccording to the following equations:

$\begin{matrix}{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}( {{{\underset{\_}{x}}_{1}h_{1}} + {{\underset{\_}{x}}_{2}h_{2}}} )h_{1}^{*}} + {\underset{\_}{n}}_{1}^{\prime}}} & (20.) \\{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}( {{{\underset{\_}{x}}_{1}{h_{1}}^{2}} + {{\underset{\_}{x}}_{2}h_{2}h_{1}^{*}}} )} + {\underset{\_}{n}}_{1}^{\prime}}} & (21.) \\{{{\underset{\_}{z}}_{1} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}}^{2} \\{h_{2}h_{1}^{*}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime}}}{and}} & (22.) \\{{\underset{\_}{z}}_{2} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}h_{2}^{*}} \\{h_{2}}^{2}\end{bmatrix}} + {\underset{\_}{n}}_{2}^{\prime}}} & (23.)\end{matrix}$

The pilot symbol sequences and x ₁ and x ₂ may be known before asequence is received and an orthogonal sequence y ^(T) may be calculatedby the loading block 214 b so that the following equations may besatisfied:y ^(H) x ₁0 and y ^(H) x ₂=0  (24.)y ^(H) z ₁=y ^(H)n₁ ^(′) and y ^(H) z ₂=y ^(H)n₂ ^(′)  (25.)In an embodiment of the invention, a unique orthogonal sequence y ^(T)may be calculated so that it is orthogonal to both pilot symbolsequences x ₁ and x ₂ at the same time. The variance of the noisecomponent n₁ ^(′) may be expressed using the following equations:

$\begin{matrix}{{{\sigma_{n_{1}^{\prime}}^{2} = {{{h_{1}}^{2}I_{oc}} = {E\lbrack {n_{1i}^{\prime}n_{1i}^{\prime*}} \rbrack}}},{i = 0},\ldots\mspace{14mu},{\frac{num\_ ded}{2} - 1}}{and}} & (26.) \\{\sigma_{n_{2}^{\prime}}^{2} = {{h_{2}}^{2}I_{oc}}} & (27.)\end{matrix}$

In instances when the orthogonal sequence y may be normalized by thedivider block 224 b, the following equation may be satisfied:y ^(H) y=1  (28.)Then the variance of y ^(H) n ₁ ^(′) may be expressed by the followingequation:E[y ^(H) n _(i) ^(′n) _(i) ^(′H) y]=σ_(n) ₁ ^(′) ²,i=1,2  (29.)Therefore, the noise power Npilot 240 b from the dedicated pilot bitsmay be obtained using the following equation:| y ^(H) z ₁|² +y ^(H) z ₂|²=σ_(n) ₁ _(′) ²+σ_(n) ₂ _(′) ²=(|h ₁|² |+|h₂|²)I _(oc)  (30.)

In instances when the number of pilot bits equals two, the two pilotbits broadcast by antenna 2 may precede the last two bits of the datafield. The pilot bits are STTD-encoded with the data and thus need to beretrieved post-STTD decoding. The receive circuitry may be adapted toextract pilot bits at the output of the combiner 206 b, post-STTDdecoding. The pilot symbol obtained post-STTD decoding may be expressedby the equation:

$\begin{matrix}{z = {{\sqrt{\frac{S_{DED}}{4}}x_{1}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}}}} & (31.)\end{matrix}$where x₁ may be the known pilot symbol sent from antenna 1 and

$\begin{matrix}{{{E\lbrack ( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} )^{2} \rbrack} = {( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} )I_{oc}}}{{pilotI} = {{{Re}(z)} = {{\sqrt{\frac{S_{DED}}{4}}I_{seq}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {{Re}( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} )}}}}} & (32.) \\{{pilotQ} = {{{Im}(z)} = {{\sqrt{\frac{S_{DED}}{4}}Q_{seq}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {{Im}( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} )}}}} & (33.)\end{matrix}$

Pilot bits pilotI and pilotQ may then be multiplied by I_(seq) andQ_(seq), respectively, to remove phase rotation. The I and Qpost-derotation components may then be used to calculate the noise powerNpilot 240 b using the following equations:σ_(n) ²=(pilotI−pilotQ)²  (34.)and σ² _(n)=(|h ₁|² +|h ₂|²)I _(oc)  (35.)In a CL1 flat fading environment, the soft value of each dedicated pilotbit at each slot on a per-finger basis may be obtained using thefollowing equations:

$\begin{matrix}{{{\underset{\_}{z}}_{1} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}}^{2} \\{{wh}_{1}^{*}h_{2}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime}}}{and}} & (36.) \\{{\underset{\_}{z}}_{2} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}}^{2} \\{{wh}_{1}^{*}h_{2}}\end{bmatrix}} + {\underset{\_}{n}}_{2}^{\prime}}} & (37.)\end{matrix}$The weight w may be calculated by firmware, so that the followingequation may be satisfied:

$\begin{matrix}{\underset{\_}{z} = {{{\underset{\_}{z}}_{1} + {w^{*}{\underset{\_}{z}}_{2}}} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{{h_{1}}^{2} + {w^{*}h_{1}h_{2}^{*}}} \\{{{wh}_{1}^{*}h_{2}} + {{w}^{2}{h_{2}}^{2}}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime} + {w^{*}{\underset{\_}{n}}_{2}^{\prime}}}}} & (38.)\end{matrix}$

The multiplying block 212 b may be used to multiply z by the orthogonalsequence y, resulting in the following equations:

$\begin{matrix}{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{\underset{\_}{y}}^{H}( {{\underset{\_}{n}}_{1}^{\prime} + {w^{*}{\underset{\_}{n}}_{2}^{\prime}}} )}} & (39.) \\{{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{\underset{\_}{y}}^{H}( {\begin{bmatrix}{n_{0}h_{1}^{\prime}} \\\vdots \\{n_{\frac{{num}\;\_\;{ded}}{2} - 1}h_{1}^{*}}\end{bmatrix} + {w^{*}\begin{bmatrix}{n_{0}h_{2}^{*}} \\\vdots \\{n_{\frac{{num}\;\_\;{ded}}{2} - 1}h_{2}^{*}}\end{bmatrix}}} )}}{and}} & (40.) \\{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{{\underset{\_}{y}}^{H}( \begin{bmatrix}\begin{matrix}{n_{0}( {h_{1}^{*} + {w^{*}h_{2}^{*}}} )} \\\vdots\end{matrix} \\{n_{\frac{{num}\;\_\;{ded}}{2} - 1}( {h_{1}^{*} + {w^{*}h_{2}^{*}}} )}\end{bmatrix} )} = {{\underset{\_}{y}}^{H}{\underset{\_}{n}}_{{cl}\; 1}}}} & (41.)\end{matrix}$The variance of the noise component n_(cl1) and determination of thenoise power Npilot 240 b may be expressed by the following equations:

$\begin{matrix}{{\sigma_{n_{{cl}\; 1}}^{2} = {{{{h_{1} + {wh}_{2}}}^{2}I_{oc}} = {E\lbrack {n_{{cl}\; 1i}n_{{cl}\; 1i}^{*}} \rbrack}}},{i = 0},\ldots\mspace{14mu},{\frac{num\_ ded}{2} - 1}} & (42.) \\{{{{\underset{\_}{y}}^{H}\underset{\_}{z}}}^{2} = \sigma_{n_{{cl}\; 1}}^{2}} & (43.)\end{matrix}$

In a CL2 flat fading environment, the same pilot pattern may be used onboth the antennas as illustrated above with regard to a CL1 environment.The soft value of each dedicated pilot bit at each slot on a per-fingerbasis may be obtained using the following equation:

$\begin{matrix}{z_{1i} = {{\sqrt{\frac{S_{DED}}{4}}( {{w_{1}h_{1}} + {w_{2}h_{2}}} )x_{i}h_{1}^{*}} + {n_{i}h_{1}^{*}}}} & (44.)\end{matrix}$The dedicated pilot symbols num_ded/2 may be stacked in a vectoraccording to the following equations:

$\begin{matrix}{{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}( {{w_{1}h_{1}} + {w_{2}h_{2}}} )\underset{\_}{x}h_{1}^{*}} + {\underset{\_}{n}}_{1}^{\prime}}}{and}} & (45.) \\{{\underset{\_}{z}}_{2} = {{\sqrt{\frac{S_{DED}}{4}}( {{w_{1}h_{1}} + {w_{2}h_{2}}} )\underset{\_}{x}\; h_{2}^{*}} + {\underset{\_}{n}}_{2}^{\prime}}} & (46.)\end{matrix}$The weights w₁ and w₂ may be calculated, using for example, firmware sothat the following equations may be satisfied:

$\begin{matrix}{\underset{\_}{z} = {{w_{1}^{*}{\underset{\_}{z}}_{1}} + {w_{2}^{*}{\underset{\_}{z}}_{2}}}} & (47.) \\{\underset{\_}{z} = {{\sqrt{\frac{S_{DED}}{4}}{{{w_{1}h_{2}} + {w_{2}h_{2}}}}^{2}\underset{\_}{x}} + {w_{1}^{*}{\underset{\_}{n}}_{1}^{\prime}} + {w_{2}^{*}{\underset{\_}{n}}_{2}^{\prime}}}} & (48.)\end{matrix}$The multiplying block 212 b may be used to multiply z by the orthogonalsequence y, which may result in the following equation:|y ^(H) z| ²=σ _(n) _(cl2) =∥w ₁ h ₁ +w ₂ h ₂∥² I _(oc)  (49.)Equation (49) may then be utilized to determine the variance of thenoise component and the noise power Npilot 240 b.

FIG. 3 is a flowchart illustrating exemplary steps for determining anoise power estimate in a WCDMA network, in accordance with anembodiment of the invention. Referring to FIGS. 2B and 3, at 302,portion of a plurality of dedicated physical channel (DPCH) pilot bitsfor a downlink channel may be summed by the summing block 206 b togenerate an in-phase (I) component 208 b and a quadrature (Q) component210 b. The generated I component and the generated Q component may bemultiplied by one or more orthogonal sequences, such as sequences 228 band 230 b, to generate at least one noise I component 232 b and at leastone noise Q component 234 b. The at least one noise I component 232 band the at least one noise Q component 234 b may be summed by thesumming block 216 b to generate at least one summed noise I component236 b and at least one summed noise Q component 238 b. The at least onesummed noise I component 236 b and the at least one summed noise Qcomponent 238 b may be squared by the squaring blocks 218 b and 220 b togenerate at least one squared noise I component and at least one squarednoise Q component. The at least one squared noise I component and the atleast one squared noise Q component may be summed by the summing block222 b to generate a resulting summed noise component. The resultingsummed noise component may be normalized by the divider block 224 b todetermine a noise power estimate Npilot 240 b for the downlink channel.

Another embodiment of the invention may provide a machine-readablestorage having stored thereon, a computer program having at least onecode section for signal processing, the at least one code section beingexecutable by a machine for causing the machine to perform steps asdisclosed herein.

Accordingly, aspects of the invention may be realized in hardware,software, firmware or a combination thereof. The invention may berealized in a centralized fashion in at least one computer system or ina distributed fashion where different elements are spread across severalinterconnected computer systems. Any kind of computer system or otherapparatus adapted for carrying out the methods described herein issuited. A typical combination of hardware, software and firmware may bea general-purpose computer system with a computer program that, whenbeing loaded and executed, controls the computer system such that itcarries out the methods described herein.

One embodiment of the present invention may be implemented as a boardlevel product, as a single chip, application specific integrated circuit(ASIC), or with varying levels integrated on a single chip with otherportions of the system as separate components. The degree of integrationof the system will primarily be determined by speed and costconsiderations. Because of the sophisticated nature of modernprocessors, it is possible to utilize a commercially availableprocessor, which may be implemented external to an ASIC implementationof the present system. Alternatively, if the processor is available asan ASIC core or logic block, then the commercially available processormay be implemented as part of an ASIC device with various functionsimplemented as firmware.

The invention may also be embedded in a computer program product, whichcomprises all the features enabling the implementation of the methodsdescribed herein, and which when loaded in a computer system is able tocarry out these methods. Computer program in the present context maymean, for example, any expression, in any language, code or notation, ofa set of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform. However, other meanings of computer program within theunderstanding of those skilled in the art are also contemplated by thepresent invention.

While the invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiments disclosed, but that the present inventionwill include all embodiments falling within the scope of the appendedclaims.

What is claimed is:
 1. A method comprising: receiving a signal on adownlink channel, the received signal having been generated based, atleast in part, on a sequence of control bits, wherein the sequence ofcontrol bits has an associated orthogonal bit sequence; calculating anoise power estimate of the received signal by extracting noisecomponents from an in-phase (I) component and a quadrature (Q) componentassociated with the received signal, wherein the noise components areextracted by multiplying the in-phase (I) component and the quadrature(Q) component by the orthogonal bit sequence; and determining asignal-to-noise-ratio (SNR) based on the calculated noise powerestimate.
 2. The method according to claim 1, further comprising:adjusting a power level based on the signal-to-noise-ratio (SNR).
 3. Themethod according to claim 1, further comprising: facilitating a handoverbased on the signal-to-noise-ratio (SNR).
 4. The method according toclaim 1, wherein said orthogonal bit sequence is generated based on aslot number of a radio frame of said received signal.
 5. The methodaccording to claim 1, wherein said orthogonal bit sequence is generatedbased on a transmit diversity mode used for transmitting said receivedsignal.
 6. The method according to claim 1, wherein the calculatingfurther comprises summing at least a portion of the control bits forsaid downlink channel to generate said in-phase (I) component and saidquadrature (Q) component.
 7. The method according to claim 6, whereinsaid control bits comprise one or more of dedicated physical channel(DPCH) transmit power control (TPC) bits, DPCH pilot bits, and commonpilot channel (CPICH) bits.
 8. The method according to claim 6, whereinthe calculating further comprises: multiplying said generated Icomponent and said generated Q component by said orthogonal bit sequenceto generate at least one noise I component and at least one noise Qcomponent; summing said at least one noise I component and said at leastone noise Q component to generate at least one summed noise I componentand at least one summed noise Q component; squaring said at least onesummed noise I component and said at least one summed noise Q componentto generate at least one squared noise I component and at least onesquared noise Q component; summing said at least one squared noise Icomponent and said at least one squared noise Q component to generate aresulting summed noise component; and normalizing said generatedresulting summed noise component to determine a noise power estimate forsaid downlink channel.
 9. The method according to claim 8, wherein saidnormalizing comprises: dividing said generated resulting summed noisecomponent by the at least the portion of control bits of said downlinkchannel.
 10. .A system for signal processing, the system comprising: areceiver configured to receive a signal on a downlink channel, thereceived signal having been generated based, at least in part, on asequence of control bits, wherein the sequence of control bits has anassociated orthogonal bit sequence; and circuitry that is configured to:calculate a noise power estimate of the received signal by beingconfigured to extract noise components from an in-phase (I) componentand a quadrature (Q) component associated with the received signal,wherein the noise components are extracted by multiplying the in-phase(I) component and the quadrature (Q) component by the orthogonal bitsequence, and determine a signal-to-noise-ratio (SNR) based on thecalculated noise power estimate.
 11. The system according to claim 10,wherein the circuitry is further configured to adjust a power levelbased on the signal-to-noise-ratio (SNR).
 12. The system according toclaim 10, wherein the circuitry is further configured to facilitate ahandover based on the signal-to-noise-ratio (SNR).
 13. The systemaccording to claim 10, wherein said orthogonal bit sequence is generatedbased on a slot number of a radio frame of said received signal.
 14. Thesystem according to claim 10, wherein said orthogonal bit sequence isgenerated based on a transmit diversity mode used for transmitting saidreceived signal.
 15. The system according to claim 10, wherein saidcircuitry enables summing at least a portion of the control bits forsaid downlink channel to generate said in-phase (I) component and saidquadrature (Q) component.
 16. The system according to claim 15, whereinsaid control bits comprise one or more of dedicated physical channel(DPCH) transmit power control (TPC) bits, DPCH pilot bits, and commonpilot channel (CPICH) bits.
 17. Tile system according to claim 15,wherein said circuitry is further configured to: multiply said generatedI component and said generated Q component by said orthogonal bitsequence to generate at least one noise I component and at least onenoise Q component; sum said at least one noise I component and said atleast one noise Q component to generate at least one summed noise Icomponent and at least one summed noise Q component; square said atleast one summed noise I component and said at least one summed noise Qcomponent to generate at least one squared noise I component and atleast one squared noise Q component; sum said at least one squared noiseI component and said at least one squared noise Q component to generatea resulting summed noise component; and normalize said generatedresulting summed noise component to determine a noise power estimate forsaid downlink channel.
 18. The system according to claim 17, whereinsaid circuitry is further configured to divide said generated resultingsummed noise component by the at least the portion of control bits ofsaid downlink channel.
 19. A system for signal processing, the systemcomprising: a receiver configured to receive a signal on an uplinkchannel, the received signal having been generated based, at least inpart, on a sequence of control bits, wherein the sequence of controlbits has an associated orthogonal bit sequence; and circuitry that isconfigured to calculate a noise power estimate of the received signal bybeing configured to extract noise components from an in-phase (I)component and a quadrature (Q) component associated with the receivedsignal, Wherein the noise components are extracted by multiplying thein-phase (I) component and the quadrature (Q) component by theorthogonal bit sequence.
 20. The system according to claim 19, whereinsaid circuitry is further configured to: determine asignal-to-noise-ratio (SNR) based on the calculated noise powerestimate.